Wireless Power Transfer System and Method

ABSTRACT

A system comprises a primary switch network coupled to a power source, wherein the primary switch network comprises a plurality of power switches, a primary resonant tank coupled to the plurality of power switches, wherein a resonant capacitor of the primary resonant tank is formed by a first variable capacitance network, and wherein the first variable capacitance network is modulated to improve soft switching of the plurality of power switches through reducing a voltage level and a current level of a switch at a turn-on instant and a primary coil coupled to the primary resonant tank.

This application claims the benefit of U.S. Provisional Application No.62/041,161, filed on Aug. 25, 2014, entitled “Cost-effective HighPerformance Wireless Power Transfer Techniques,” which application ishereby incorporated herein by reference.

TECHNICAL FIELD

The present invention relates to high performance power supplies, and,in particular embodiments, to a wireless power transfer system.

BACKGROUND

As technologies further advance, wireless power transfer (WPT) hasemerged as an efficient and convenient mechanism for powering orcharging battery based mobile devices such as mobile phones, tablet PCs,digital cameras, MP3 players and/or the like. A wireless power transfersystem typically comprises a primary side transmitter and a secondaryside receiver. The primary side transmitter is magnetically coupled tothe secondary side receiver through a magnetic coupling. The magneticcoupling may be implemented as a loosely coupled transformer having aprimary side coil formed in the primary side transmitter and a secondaryside coil formed in the secondary side receiver.

FIG. 1 illustrates a block diagram of a wireless power transfer system.The wireless power transfer system shown in FIG. 1 is an exemplarysystem required by the Alliance for Wireless Power (A4WP) organization.The wireless power transfer system shown in FIG. 1 includes a powertransmitter and a power receiver. Through magnetic coupling, power istransferred from the power transmitter to the power receiver.

The power transmitter includes a transmitter dc/dc converter, a poweramplifier, an impedance matching circuit and a resonant circuitconnected in cascade between a power input and a transmitter coil. Thepower transmitter further comprises a transmitter Bluetooth unit havinga first input/output coupled to a receiver Bluetooth unit and a secondinput/output coupled to the transmitter dc/dc converter of the powertransmitter. The power receiver includes a resonant circuit, arectifier, a receiver dc/dc converter connected in cascade between areceiver coil and a load. The power receiver further comprises thereceiver Bluetooth unit having a second input/input coupled to thereceiver dc/dc converter.

According to the standard of A4WP, the power transmitter operates at afixed system frequency within a frequency band ranging from 6.765 MHz to6.795 MHz (nominal 6.78 MHz). The transmitter converts dc power at itsinput to high frequency ac power within the frequency band. Thetransmitter coil, coupled to the power amplifier through a resonantcircuit (usually one or more capacitors), forms a transmitter resonanttank with the resonant circuit and generates a magnetic field at thesystem frequency. Through magnetic coupling, power is transferred to thereceiver coil nearby. Likewise, the receiver coil and the resonantcircuit of the power receiver form a receiver resonant tank.

Both the resonant circuit coupled to the receiver coil and the resonantcircuit coupled to the transmitter coil may comprise one or morecapacitors. The resonant frequency of the transmitter resonant tank andthat of the receiver resonant tank are designed to be at the systemfrequency, which is determined by the switching frequency of the poweramplifier.

In order to match the power capability and electrical parameters of thepower amplifier and those of the resonant tank in the power transmitter,an impedance matching circuit is coupled between the power amplifier andthe transmitter resonant circuit as shown in FIG. 1.

The rectifier in the power receiver converts high frequency ac powerfrom the receiver coil into dc power and delivers the dc power to theload through the receiver dc/dc converter. In the system shown in FIG.1, for a given input voltage Vin sent to the power amplifier, the outputvoltage Vo at the rectifier may vary in a wide range due to a variety offactors such as the coupling coefficient changes between the transmitterand the receiver, load changes and the like. In order to control theoutput voltage within an acceptable range, the transmitter dc/dcconverter may be employed to control the voltage sent to the poweramplifier, and the receiver dc/dc converter may be employed to furtherregulate the voltage fed to the load. Because the input power is mostlikely from an ac/dc adapter plugged into an ac source, the transmitterdc/dc converter is implemented as a dc/dc converter. Similarly, thereceiver dc/dc converter is usually implemented as a dc/dc converter.The load can be actual loads such as integrated circuits (ICs), abattery and the like. Alternatively, the load can be a downstreamconverter such as a battery charger, a dc/dc converter coupled to anactual load and the like.

The transmitter Bluetooth unit and the receiver Bluetooth unit form aBluetooth communication subsystem providing a communication channelbetween the power receiver and the power transmitter. For example, thevoltage control signal may be communicated through this Bluetoothcommunication subsystem. It should be noted that other communicationtechniques such as WiFi, Zigbee devices and the like, can also be usedfor the communication between the power transmitter and the powerreceiver. Furthermore, an in-band communication between the powerreceiver and the power transmitter can also be implemented by modulatingthe power signal transferred between the power receiver and the powertransmitter.

The system shown in FIG. 1 includes many stages. Many components in thesystem shown in FIG. 1 may have high voltage/current stresses. As such,the system shown in FIG. 1 is a complex system, which causes high costand low efficiency.

SUMMARY OF THE INVENTION

These and other problems are generally solved or circumvented, andtechnical advantages are generally achieved, by preferred embodiments ofthe present invention which improve efficiency of a wireless powertransfer system.

In accordance with an embodiment, a system comprises a primary switchnetwork coupled to a power source, wherein the primary switch networkcomprises a plurality of power switches, a primary resonant tank coupledto the plurality of power switches, wherein a resonant capacitor of theprimary resonant tank is formed by a first variable capacitance network,and wherein the first variable capacitance network is modulated toimprove soft switching of the plurality of power switches throughreducing a voltage level and a current level of a switch at a turn-oninstant and a primary coil coupled to the primary resonant tank.

In accordance with another embodiment, a method comprises detecting asignal representing a current level at a turn-on transition of a powerswitch of a primary switch network, wherein the primary switch networkis coupled to a primary resonant tank having a resonant capacitor formedby a first variable capacitance network comprising a plurality ofswitch-capacitor networks and modulating a capacitance of the firstvariable capacitance network to improve soft switching of the powerswitch through reducing a voltage level and a current level at theturn-on transition of the power switch.

In accordance with yet another embodiment, a method comprises providinga wireless power transfer system comprising a transmitter magneticallycoupled to a first receiver, wherein the transmitter comprises a poweramplifier coupled to an input power source, a transmitter resonant tankcomprising a first variable capacitance network and a transmitter coilcoupled to the transmitter resonant tank and the first receivercomprises a first receiver resonant tank comprising a first receivervariable capacitance network and a first receiver coil coupled to thefirst receiver resonant tank.

The method further comprises modulating a capacitance of the firstvariable capacitance network so that switches of the power amplifierachieve an improved soft switching condition.

An advantage of a preferred embodiment of the present invention isimproving a wireless power transfer system's performance throughadjusting at least one resonant component of the wireless power transfersystem.

The foregoing has outlined rather broadly the features and technicaladvantages of the present invention in order that the detaileddescription of the invention that follows may be better understood.Additional features and advantages of the invention will be describedhereinafter which form the subject of the claims of the invention. Itshould be appreciated by those skilled in the art that the conceptionand specific embodiment disclosed may be readily utilized as a basis formodifying or designing other structures or processes for carrying outthe same purposes of the present invention. It should also be realizedby those skilled in the art that such equivalent constructions do notdepart from the spirit and scope of the invention as set forth in theappended claims.

BRIEF DESCRIPTION OF THE DRAWINGS

For a more complete understanding of the present invention, and theadvantages thereof, reference is now made to the following descriptionstaken in conjunction with the accompanying drawings, in which:

FIG. 1 illustrates a block diagram of a wireless power transfer system;

FIG. 2 illustrates a block diagram of a first implementation of a powertransmitter of a wireless power transfer system in accordance withvarious embodiments of the present disclosure;

FIG. 3 illustrates a block diagram of a wireless power transfer systemin accordance with various embodiments of the present disclosure;

FIG. 4 illustrates a schematic diagram of a first illustrativeimplementation of the first EMI filter of the wireless power transfersystem shown in FIG. 3 in accordance with various embodiments of thepresent disclosure.

FIG. 5 illustrates a schematic diagram of a first illustrativeimplementation of the harmonic notch circuit shown in FIG. 4 inaccordance with various embodiments of the present disclosure;

FIG. 6 illustrates a schematic diagram of a wireless power transfersystem having an impedance matching circuit in accordance with variousembodiments of the present disclosure;

FIG. 7 illustrates a variety of waveforms associated with the wirelesspower transfer system shown in FIG. 6 in accordance with variousembodiments of the present disclosure;

FIG. 8 illustrates a block diagram of a wireless power transfer systemwith better efficiency on the basis of controlled resonance enabled by avariable capacitance technique in accordance with various embodiments ofthe present disclosure;

FIG. 9 illustrates a block diagram of a wireless power transfer systemcontrolled by a resonant component modulation technique in accordancewith various embodiments of the present disclosure;

FIG. 10 illustrates a schematic diagram of the wireless power transfersystem shown in

FIG. 9 in accordance with various embodiments of the present disclosure;

FIG. 11 illustrates a block diagram of an implementation of a feedbackcontrol system of a wireless power transfer system in accordance withvarious embodiments of the present disclosure;

FIG. 12 shows a variety of waveforms associated with a wireless powertransfer system with weak coupling between the transmitter and thereceiver coils in accordance with various embodiments of the presentdisclosure;

FIG. 13 shows a variety of waveforms associated with a wireless powertransfer system with the same coupling as that of FIG. 12 in accordancewith various embodiments of the present disclosure;

FIG. 14 shows a variety of waveforms associated with a wireless powertransfer system with stronger coupling between the transmitter and thereceiver coils in accordance with various embodiments of the presentdisclosure;

FIG. 15 illustrates a schematic diagram of a variable capacitancenetwork in accordance with various embodiments of the presentdisclosure;

FIG. 16 illustrates a schematic diagram of a zero-voltage switchingasymmetric half-bridge converter in accordance with various embodimentsof the present disclosure;

FIG. 17 illustrates a variety of waveforms associated with thezero-voltage switching asymmetric half-bridge converter shown in FIG. 16in accordance with various embodiments of the present disclosure;

FIG. 18 illustrates a variety of waveforms associated with thezero-voltage switching asymmetric half-bridge converter operating in theultra-light load mode in accordance with various embodiments of thepresent disclosure; and

FIG. 19 illustrates a cross sectional view of an integrated magneticstructure employed in FIG. 16 in accordance with various embodiments ofthe present disclosure.

Corresponding numerals and symbols in the different figures generallyrefer to corresponding parts unless otherwise indicated. The figures aredrawn to clearly illustrate the relevant aspects of the variousembodiments and are not necessarily drawn to scale.

DETAILED DESCRIPTION OF ILLUSTRATIVE EMBODIMENTS

The making and using of the presently preferred embodiments arediscussed in detail below. It should be appreciated, however, that thepresent invention provides many applicable inventive concepts that canbe embodied in a wide variety of specific contexts. The specificembodiments discussed are merely illustrative of specific ways to makeand use the invention, and do not limit the scope of the invention.

The present invention will be described with respect to preferredembodiments in a specific context, namely a wireless power transfersystem having a plurality of variable capacitance networks. Theinvention may also be applied, however, to a variety of other powersystems. Hereinafter, various embodiments will be explained in detailwith reference to the accompanying drawings.

FIG. 2 illustrates a block diagram of a first implementation of a powertransmitter of a wireless power transfer system in accordance withvarious embodiments of the present disclosure. The power transmitter 200includes a power converter 202, a power amplifier 204, an optionalimpedance matching circuit 206 and a resonant circuit 208 connected incascade between a power input Vin and a transmitter coil. The powertransmitter 200 further comprises a communication device 212 and afrequency generation unit 214. A reference clock generated by thecommunication device 212 is fed into the frequency generation unit 214.The frequency generation unit 214 generates a system frequency signalfed into the power amplifier 204 as shown in FIG. 2.

To meet stringent EMC requirements of wireless power transfer systems,the power amplifier 204 of the power transmitter 200 may be required toprovide a current or voltage in a sinusoidal shape. Such a current orvoltage in a sinusoidal shape fed into the transmitter coil is requiredto be within an Industrial, Scientific and Medical (ISM) frequency band.In order to maintain the voltage or current signal generated by thepower amplifier 204 within the ISM band, high-accuracy components areneeded to generate a system frequency signal.

FIG. 2 illustrates a mechanism of generating a system frequency signalbased upon a frequency signal in the communication system. In wirelesspower transfer systems, there may be several clocks available. Forexample, a Bluetooth device may have a plurality of system clocks suchas a 3.2 kHz native clock, a reference clock, other high accuracyderived clocks, and RF clocks which correspond to the RF carrierfrequencies of its physical RF channels. Both system clocks and RFclocks can be used to generate a reference signal Fr as shown in FIG. 2.In some embodiments, the system clocks have a lower frequency. Such alower frequency signal is able to travel a longer distance withoutcausing noise issues. As such, the system clocks may be employed as areference frequency for the power amplifier 204. The system frequency Fsshown in FIG. 2 can be generated by using simple circuits such ascounter-based frequency multipliers, frequency dividers and the like.The frequency generation unit 214 may comprise the simple circuitsdescribed above and generate the system frequency signal Fs having afrequency within a frequency band to which it is specified.

In some embodiments, the system frequency Fs has a frequency k times thereference frequency Fr, wherein k is an integer. In alternativeembodiments, the system frequency Fs has a frequency equal to thereference frequency Fr divided by k. In some embodiments, the systemfrequency Fs is sent to the power amplifier 204 and used to control theswitches of the power amplifier 204.

In some embodiments, the frequency generation unit 214 is part of acommunication device. Other control and protection functions of thepower transmitter may be implemented through a microcontroller, a statemachine or other circuits, and may be integrated with the Bluetoothfunction into one IC. In alternative embodiments, the frequencygeneration unit 214 is part of the power amplifier 204. Furthermore, thefrequency generation unit 214 may be implemented as an independent partcoupled between the communication device 212 and the power amplifier204.

Radiated RF emission is a very important concern for EMC compliance in awireless power transfer system. One important consideration is to reducethe interference caused by the currents in the transmitter coil and thereceiver coil. For this purpose, EMI filters are employed to reduce thehigh frequency components of the currents in the transmitter coil aswell as the high frequency components of the currents in the receivercoil.

FIG. 3 illustrates a block diagram of a wireless power transfer systemin accordance with various embodiments of the present disclosure. Theblock diagram of the wireless power transfer system 300 shown in FIG. 3is similar to that shown in FIG. 1 except that a first EMI filter 302 iscoupled between the power amplifier and the resonant circuit of thepower transmitter, and a second EMI filter 312 is coupled between theresonant circuit and the rectifier of the power receiver. Forsimplicity, only the first EMI filter 302 and the second EMI filter 312are described in detail herein. It should be noted that the impedancematching circuit shown in FIG. 1 can be placed before or after the firstEMI filter 302. Furthermore, the impedance matching circuit may be partof the first EMI filter 302.

In some applications, it's also feasible to have an EMI filter only inthe power transmitter, or only in the power receiver. In someembodiments, the EMI filters shown in FIG. 3 may have differentconfigurations such as low-pass filters, band-pass filters and othersuitable topologies. In some embodiments, the EMI filters shown in FIG.3 may comprise inductors and capacitors which form resonant circuitswith one or more resonant frequencies. The detailed structures of theEMI filters will be described below with respect to FIGS. 4-5.

FIG. 4 illustrates a schematic diagram of a first illustrativeimplementation of the first EMI filter of the wireless power transfersystem shown in FIG. 3 in accordance with various embodiments of thepresent disclosure. FIG. 4 shows a power transmitter is coupled to apower receiver. A detailed schematic diagram of the power receiver isnot shown for the sake of brevity. The power transmitter comprises apower amplifier 402, an EMI filter 404 and a resonant circuit comprisingCrt connected in series between Vin and the transmitter coil Lt.

In some embodiments, the power amplifier 402 is implemented as a class-Dpower amplifier as shown in FIG. 4. The power amplifier 402 comprisesswitches S1 and S2 connected in series between two terminals of Vin. Thecommon node of switches S1 and S2 is connected to an input of the EMIfilter 404. It should be noted that the impedance matching circuit isnot shown. Depending on different applications and design needs, theimpedance matching circuit can be placed before or after the EMI filter404.

The EMI filter 404 comprises inductors L1, L2, L3 and L4, and capacitorsC1, C2, C3 and C4. As shown in FIG. 4, L1 and C1 are connected inparallel. L2 and C2 are connected in parallel. L3 and C3 are connectedin series. L4 and C4 are connected in series. In some embodiments, L1and C1 form a first harmonic trap circuit; L2 and C2 form a secondharmonic trap circuit; L3 and C3 form a first harmonic notch circuit; L4and C4 form a second harmonic notch circuit.

It should be noted that FIG. 4 illustrates only one capacitor isincluded in each harmonic suppression circuit. This is merely anexample. Each harmonic suppression circuit shown in FIG. 4 may includehundreds of such capacitors. The number of capacitors illustrated hereinis limited solely for the purpose of clearly illustrating the inventiveaspects of the various embodiments. The present invention is not limitedto any specific number of capacitors.

The resonant frequencies of the harmonic trap circuits and harmonicnotch circuits can be set to the frequencies at which harmonics will besuppressed. In some embodiments, a harmonic trap circuit and acorresponding harmonic notch circuit can have the same resonantfrequency. For example, in FIG. 4, both the first harmonic trap circuitcomprising L1 and C1, and the first harmonic notch circuit comprising ofL3 and C3 can be designed for suppressing the 3rd harmonic. Since the3rd harmonic is a dominant harmonic, it needs more filtering than otherhigher order harmonics. Using both the first harmonic trap circuit andthe first harmonic notch circuit to suppress the 3^(rd) harmonic helpsthe EMI filter achieve better harmonic suppression.

In some embodiments, the second harmonic trap circuit comprising L2 andC2 can be set for suppressing the 5th harmonic. The second harmonicnotch circuit comprising of L4 and C4 can be set for suppressing the 7thharmonic. As such, the 3rd, 5th and 7th harmonic currents will bereduced significantly, and other higher order harmonics can also besuppressed. As a result, the current in the transmitter coil Lt will besubstantially sinusoidal. It should be noted that in order to achievebetter system performance, it is desirable to have a low inductance pathin any harmonic trap circuit shown in FIG. 4 and a low capacitance pathin any harmonic notch circuit shown in FIG. 4. Such a configurationhelps to reduce the impact on the wireless power transfer systemoperating at the fundamental frequency. It should be noted that thefundamental frequency is equal to or approximately equal to the systemfrequency of the wireless power transfer system.

In practical filter implementations, it is important to make sure thatkey resonant frequencies of the EMI filter match the frequencies towhich they are specified. For example, it is desirable that the resonantfrequencies of the harmonic trap circuits and the harmonic notchcircuits are close to the desired values. In order to achieve this withminimum costs and efforts, some inductors and capacitors shown in FIG. 4can be integrated into one package using suitable semiconductorfabrication processes. The values of inductors and/or capacitors can betrimmed in the fabrication processes so as to achieve the requiredresonant frequencies. The detailed trimming process will be describedbelow with respect to FIG. 5.

FIG. 5 illustrates a schematic diagram of a first illustrativeimplementation of the harmonic notch circuit shown in FIG. 4 inaccordance with various embodiments of the present disclosure. Theharmonic notch circuit shown in FIG. 4 comprises L3 and C3. In the firstimplementation of the harmonic notch circuit shown in FIG. 5, C3 may bereplaced by a plurality of capacitors. As shown in FIG. 5, C3 is atrimmable capacitor including C10, C11, C12, C13 and C14.

In some embodiments, both L3 and C3 may be fabricated on a samesubstrate. The capacitor C3 is manufactured onto a semiconductorsubstrate using a first semiconductor fabrication process. The inductorL3 is manufactured onto the semiconductor substrate comprising thecapacitor C3. In alternative embodiments, L3 is manufactured onto aseparate semiconductor substrate, and then stacked on the semiconductorsubstrate comprising the capacitor C3. Furthermore, L3 is manufacturedonto a separate substrate coupled to the semiconductor substratecomprising the capacitor C3. Furthermore, L3 may be a discrete componentcoupled to the package comprising the capacitor C3. The fabricationprocesses above are well known in the art, and hence are not discussedin further detail herein.

In some embodiments, all capacitors shown in FIG. 5 are capacitorsfabricated on a semiconductor substrate. In alternative embodiments, atleast one capacitor (e.g., capacitor C10) is a discrete capacitorcoupled to the semiconductor substrate on which the other capacitors arefabricated. The capacitance of the discrete capacitor is greater thanthe total capacitance of the capacitors fabricated on the semiconductorsubstrate.

As shown in FIG. 5, capacitor C10 and inductor L3 are connected inseries. Capacitor C11 and a first trim device F11 are connected inseries and further connected in parallel with capacitor C10. Likewise,capacitor C12 and a second trim device F12 are connected in series andfurther connected in parallel with capacitor C10; capacitor C13 and athird trim device F13 are connected in series and further connected inparallel with capacitor C10; capacitor C14 and a fourth trim device F14are connected in series and further connected in parallel with capacitorC10. It should be recognized that while FIG. 5 illustrates the trimmablecapacitor C3 comprising four trim devices and their correspondingcapacitors, the trimmable capacitor C3 could accommodate any number oftrim devices and their corresponding capacitors.

During the fabrication process of the trimmable capacitor C3, each trimdevice is initially in a short circuit condition. Depending on designneeds, the trim device shown in FIG. 5 may be turned into an opencircuit so as to change the total capacitance of the trimmable capacitorC3. In some embodiments, the trim device shown in FIG. 5 may beimplemented as any suitable semiconductor elements such as a metaltrace, a fuse, a low-value resistor or any similar components having avalue change from a short-circuit (low resistance) state to anopen-circuit (high resistance) state by applying electrical energy to itor through a mechanical force such as laser cutting.

During the fabrication process, a variety of factors may have an impacton the final values of L3 and C10. Furthermore, the interconnectcomponents coupled between L3 and C10 may result in further inaccuraciesof the resonant frequency of the LC resonant network. By changing thestate of the trim devices, the accuracy of the resonant frequency may beimproved by selecting the number of capacitors connected in parallelwith C10.

In some embodiments, the capacitors shown in FIG. 5 follow a binaryrelationship so as to simplify the trim process. In particular, thecapacitance of C11 is equal to one half of the capacitance of C10; thecapacitance of C12 is equal to one half of the capacitance of C11; thecapacitance of C13 is equal to one half of the capacitance of C12; thecapacitance of C14 is equal to one half of the capacitance of C13.

The values of these trim devices can be decided in the manufacturingprocess by assessing the actual values of the capacitor C3 and theinductor L3, or by assessing the impedance of the LC network. Forexample, after L3 and C3 have been manufactured onto the package, theimpedance between point a and point b shown in FIG. 5 can be tested withall capacitors C10 through C14 connected in parallel to find theresonant frequency of the LC series resonant circuit. By comparing themeasured resonant frequency with the desired resonant frequency, it ispossible to figure out the percentage of capacitance to be trimmed out.Then the corresponding trim devices can be turned into an open-circuitstate to get the appropriate capacitance for C3.

It should be noted that the trimming process described above withrespect to FIG. 5 is merely an example. A person skilled in the artwould understand the trimming process may be applicable to other EMIelements shown in FIG. 4.

A complex LC network such as a plurality of the harmonic trap circuitsand harmonic notch circuits, or even the EMI filters shown in FIG. 4,can be integrated into a package with the help of semiconductormanufacturing process in a similar way. To get the desired resonantfrequencies, a plurality of capacitors can be trimmed in a mannersimilar to that shown in FIG. 5.

It should be noted that when a trimming process is applicable tomultiple branches in parallel, it is important that the parallelconnection should be not connected inside the package. For example, ifC3 and/or C4 of the EMI filter shown in FIG. 4 need to be trimmed, pointA and point B shown in FIG. 4 should be connected individually to twoseparate interconnection pins of the package, but not shorted inside thepackage. By connecting these points outside the package, the value ofeach component as well as the impedance of each branch (e.g., onecomprising of L3 and C3, and another comprising of L4 and C4) can bemeasured correctly. In some embodiments, these two interconnection pinscan be connected together by a metal trace later on a system board. Inthis way, the filter functions will be performed at the system level,while parallel branches of the filter elements can be trimmedindividually since these two branches are separated from each otherduring the trimming process.

It should be note that the EMI filter topologies shown in FIGS. 4-5 aremerely examples, which should not unduly limit the scope of the claims.One of ordinary skill in the art would recognize many variations,alternatives, and modifications. It should further be noted that all orpart of the EMI filters shown in FIGS. 4-5 may be integrated with thepower amplifier and/or the coils by using suitable fabricationprocesses.

FIG. 6 illustrates a schematic diagram of a wireless power transfersystem having an impedance matching circuit in accordance with variousembodiments of the present disclosure. The wireless power transfersystem 600 comprises a transmitter 602 and a receiver 612. Thetransmitter 602 comprises a power amplifier 604 formed by S1 and S2connected in series, an impedance matching circuit 606, a transmitterresonant circuit formed by Crt coupled between Vin and a transmittercoil Lt. The receiver 612 comprises a receiver resonant circuit formedby Crr, a rectifier formed by D1 and D2 and an output capacitor Cocoupled between a receiver coil Lr and a load.

The impedance matching circuit 606 comprises a first inductor L1, asecond inductor L2 and a first capacitor C1. As shown in FIG. 6, thefirst inductor L1 and the second inductor L2 are connected in series.The first capacitor C1 is connected to the common node of the firstinductor L1 and the second inductor L2.

In some embodiments, the values of L1, L2 and C1 are calculated basedupon the impedance matching requirements of the wireless power transfersystem 600. However, if these components are chosen just based on thecalculated values, S1 and S2 may not achieve soft-switching. As aresult, the power losses in the power amplifier may be excessively high.In some embodiments, the value of C1 may be adjusted to a value slightlyaway from the calculated value of C1. By adjusting the value of C1, thepower amplifier 604 may achieve soft-switching with a minimum impact onthe impedance matching function of the impedance matching circuit 606.

FIG. 7 illustrates a variety of waveforms associated with the wirelesspower transfer system shown in FIG. 6 in accordance with variousembodiments of the present disclosure. The horizontal axis of FIG. 7represents intervals of time. The unit of the horizontal axis isnanosecond. There may be four vertical axes. The first vertical axis Y1represents the voltage (V_(SW)) across the drain-to-source of the switchS2, the gate drive voltage (V_(s2g)) of the switch S2 and the outputvoltage Vo. The second vertical axis Y2 represents the current (I_(SW))flowing through the first inductor L1 and the voltage (V_(C1)) acrossthe first capacitor C1. The third vertical axis Y3 represents thecurrent (I_(t)) flowing through the transmitter coil Lt and the voltage(V_(crt)) across the capacitor Crt. The fourth vertical axis Y4represents the current (I_(r)) flowing through the receiver coil Lr andthe voltage (V_(crr)) across the capacitor Crr.

From the waveform of Vsw, it is clear that the power switches (e.g.,power switch S2) are turned on with a voltage approximately equal tozero. As a result, soft-switching is achieved. Vs2 g is the gate drivevoltage of S2. As shown in FIG. 7, Vs2 g starts to rise after Vsw isreduced to a voltage approximately equal to zero. It should be notedthat the soft switching described above is achieved through adjustingthe capacitance of C1 in response to different operating conditions suchas different output power and/or different input voltages. Furthermore,the adjustment of the capacitance of C1 is also related to the changesof the inductance of the transmitter coil, the inductance of thereceiver coil and/or the coupling between the transmitter coil and thereceiver coil.

To maintain soft-switching over a wide range of operating conditionswithout causing too much current and voltage stresses on the powercomponents in the transmitter, it is better to change the capacitance ofC1 adaptively. U.S. patent application Ser. No. 14/177,049 entitled“High Efficiency High Frequency Resonant Power Conversion” by the sameinventor of the present application discloses techniques to change thecapacitance adaptively. Such techniques can be applicable to the presentapplication so as to change the capacitance of C1 in real time. In otherwords, the capacitance of C1 can be adjusted or modulated dynamicallyaccording to different system needs.

FIG. 8 illustrates a block diagram of a wireless power transfer systemwith better efficiency on the basis of controlled resonance enabled by avariable capacitance technique in accordance with various embodiments ofthe present disclosure. The structure of the wireless power transfersystem 800 shown in FIG. 8 is similar to that shown in FIG. 3 exceptthat a transmitter controller 802 and a receiver controller 812 areemployed to adjust the capacitances of the transmitter resonant circuitand the receiver resonant circuit respectively.

The transmitter controller 802 takes information from the input powerconverter, the power amplifier, the transmitter resonant tank and theBluetooth communication unit in the transmitter. Based upon theinformation, the transmitter controller 802 is capable of adjusting theoperation of the power transmitter by modulating the capacitance and/orinductance in the transmitter resonant circuit coupled to thetransmitter coil. Furthermore, the transmitter controller 802 is able todynamically change the timing of the switches in the power amplifier andchange the input voltage Vin fed into the power amplifier.

The receiver controller 812 takes information from the output powerconverter, the output voltage of the rectifier Vo, the receiver resonantcircuit and the Bluetooth communication unit in the receiver. Based uponthe information, the receiver controller is capable of adjusting theoperation of the receiver by modulating the capacitance and/orinductance in the receiver resonant tank coupled to the receiver coil.

In this way, the local controllers (e.g., the transmitter controller 802and the receiver controller 812) provide fast control actions, while theBluetooth communication units can provide slow control and adjustmentfunctions. It should be noted that impedance matching circuit isoptional to the system shown in FIG. 8 and may not be needed during themajority of time, so it is not shown in FIG. 8 for the sake of brevity.

One advantageous feature of having the capacitance modulation techniquedescribed above is that the resonant frequencies of the transmitterresonant circuit and the receiver resonant circuit can be dynamicallyfine-tuned so that the resonant frequencies are equal to orapproximately equal to the system frequency of the wireless powertransfer system (e.g., 6.78 MHz in an A4WP based system). As a result,the efficiency of the wireless power transfer system 800 is improved.

Another advantageous feature of having the capacitance modulationtechnique described above is the power processing including the powertransfer between the power transmitter and the power receiver may beimproved. For example, the output voltage Vo may be regulated throughmodulating the capacitances. Such a regulated output voltage helps tosave the output power converter and/or the input power converter shownin FIG. 8. As a result, the system cost may be reduced. The detailedimplementation of this advantageous feature will be described below withrespect to FIGS. 9-15.

FIG. 9 illustrates a block diagram of a wireless power transfer systemcontrolled by a resonant component modulation technique in accordancewith various embodiments of the present disclosure. The wireless powertransfer system 900 is similar to the system shown in FIG. 8 except thatboth the input power converter and the output power converter shown inFIG. 8 have been removed as a result of applying the resonant componentmodulation technique to the wireless power transfer system 900.

In the wireless power transfer system 900, the power control, the outputvoltage regulation and the soft-switching operation for the powerswitches are achieved by modulating the resonant component values in thetransmitter resonant circuit and the receiver resonant circuit. Theresonance modulation technique is employed to adjust the impedance ofthe resonant tanks dynamically, thereby controlling both the reactivepower and active power of the wireless power transfer system 900.

It is important to regulate both the active power and reactive power tomaintain an optimum operation of the system. Especially, the impedanceof the power receiver presented to the power transmitter can be adjustedby modulating the impedance of the power receiver. Thus, the modulationof the impedance of the power receiver plays a role similar to thefunction of an impedance matching circuit. In other words, the impedancematching circuit (not shown) in the receiver may be replaced bymodulating the impedance of the power receiver. As a result, it is notnecessary to have a separate impedance matching circuit in the powerreceiver and/or the power transmitter.

FIG. 10 illustrates a schematic diagram of the wireless power transfersystem shown in FIG. 9 in accordance with various embodiments of thepresent disclosure. The wireless power transfer system 1000 comprises apower transmitter 1002 and a power receiver 1012 coupled togetherthrough magnetic coupling. The strength of coupling between the powertransmitter 1002 and the power receiver 1012 is quantified by thecoupling coefficient k. In some embodiments, k is in a range from about0.05 to about 0.9. Although only one receiver is shown in FIG. 10,multiple receivers may be coupled to the power transmitter 1002.

The power transmitter 1002 comprises a power amplifier 1004, atransmitter EMI filter 1006, a transmitter resonant circuit 1008connected in series between Vin and a transmitter coil Lt. The poweramplifier 1004 is implemented as a class D power amplifier comprisingswitches S1 and S2. The power amplifier 1004 shown in FIG. 10 is avoltage-fed half-bridge topology. It should be noted that the powertopology of the power amplifier 1004 shown in FIG. 10 is merely anexample. A person skilled in the art will recognize there may be manyalternatives, variations and modifications. For example, other suitablevoltage-fed topologies such as full-bridge converters, push-pullconverters could be employed. Furthermore, current-fed topologies suchas class-E and current-fed push-pull topologies may also be used.

The transmitter EMI filter 1006 comprises inductors L1, L2, L3 and L4,and capacitors C1, C2, C3 and C4. As shown in FIG. 10, L1 and C1 areconnected in parallel. L2 and C2 are connected in parallel. L3 and C3are connected in series. L4 and C4 are connected in series. In someembodiments, L1 and C1 form a first harmonic trap circuit; L2 and C2form a second harmonic trap circuit; L3 and C3 form a first harmonicnotch circuit; L4 and C4 form a second harmonic notch circuit.

The transmitter resonant circuit 1008 comprises a resonant capacitorCrt. Crt can be implemented as a capacitor having variable capacitance.For example, Crt may be implemented as a capacitor and switch network asdescribed in U.S. patent application Ser. No. 14/177,049. Thecapacitance of Crt can be modulated by controlling the gate signalsapplied to the switches in the capacitor and switch networks accordingto different system operating conditions. The arrangement of thecapacitors and switches in the capacitor and switch networks is designedsuch that the capacitor and switch networks are capable of generating alarge number of capacitance variation steps, which offer an almostcontinuous variation of the capacitance of Crt in a wide range.

The power receiver 1012 comprises a rectifier 1014, a receiver EMIfilter 1016, a receiver resonant circuit 1018 connected in seriesbetween a load and a receiver coil Lr. The receiver EMI filter 1016comprises inductors L5, L6, and L7, and capacitors C5, C6, and C7. Asshown in FIG. 10, L5 and C5 are connected in parallel. L6 and C6 areconnected in series. L7 and C7 are connected in series. In someembodiments, L5 and C5 form a harmonic trap circuit in the receiver EMIfilter 1016; L6 and C6 form a first harmonic notch circuit in thereceiver EMI filter 1016; L7 and C7 form a second harmonic notch circuitin the receiver EMI filter 1016.

The rectifier 1014 comprises diodes D1 and D2. In alternativeembodiments, D1 and D2 can also be implemented as synchronousrectifiers. For example, MOSFETs are controlled to emulate diodefunctions. Furthermore, the rectifier 1014 may be formed by other typesof controllable devices such as bipolar junction transistor (BJT)devices, super junction transistor (SJT) devices, insulated gate bipolartransistor (IGBT) devices, gallium nitride (GaN) based power devicesand/or the like. The detailed operation and structure of the rectifier1014 are well known in the art, and hence are not discussed herein.

The receiver resonant circuit 1018 comprises a resonant capacitor Crr.Crt can be implemented as a capacitor having variable capacitance. Forexample, Crr may be implemented as a capacitor and switch network asdescribed in U.S. patent application Ser. No. 14/177,049. Thecapacitance of Crr can be modulated by controlling the gate signalsapplied to the switches in the capacitor and switch networks accordingto different system operating conditions. The arrangement of thecapacitors and switches in the capacitor and switch networks is designedsuch that the capacitor and switch networks are capable of generating alarge number of capacitance variation steps, which offer an almostcontinuous variation of the capacitance of Crr in a wide range.

Lt and Lr are the transmitter coil of the transmitter 1002 and thereceiver coil of the receiver 1012 respectively. In operation, Lt and Lrare placed in proximity physically, so their magnetic fields are coupledtogether. The coupling between Lt and Lr depends on the relativeposition and orientation of these two coils, and thus may vary in a widerange in a practical wireless power transfer system.

The load can be actual loads such as integrated circuits (ICs), abattery and the like. Alternatively, the load can be a downstreamconverter such as a battery charger, a dc/dc converter coupled to anactual load and the like.

In some embodiments, the output voltage Vo is a regulated voltage. Inalternative embodiments, the output voltage Vo is maintained within arange to which it is specified. The output power of the wireless powersystem 1000 is equal to the output voltage Vo times the output currentIo. In some embodiments, the output current Io as well as the outputpower Po may vary in a wide range depending on different operatingcondition, while the output voltage Vo is either a regulated voltage orwithin a narrow range (e.g., +/−10% of the regulated voltage).

As described above with respect to FIG. 10, both Crt and Crr can bemodulated in response to different system operating conditions. As such,there may be two control variables derived from modulating Crt and Crr.One control variable may be used to control the output voltage as wellas the output power. The other control variable may be used to improvethe performance of the wireless power transfer system such as improvingefficiency, reducing voltage and current stresses, maximizing the powertransfer and/or the like.

The performance improvement may be realized through improving theefficiency of the wireless power transfer system. Since the wirelesspower transfer system is a complex system, it is a slow and difficultprocess to improve the efficiency of such a complex system. In someembodiments, better efficiency may be achieved by operating the poweramplifier based upon the following two principles.

In accordance with a first principle of achieving soft switching, bothS1 and S2 can achieve soft switching if S1 and S2 are turned on when thevoltages across S1 and S2 are equal to or approximately equal to zero.In accordance with a second principle, the efficiency of the wirelesspower transfer system may be further improved when the current Isw ofthe power amplifier 1004 is kept low during the turn-on transitions ofS1 and S2 because a lower Isw helps to reduce the switching losses aswell as the conduction losses of S1 and S2. In other words, in order toachieve a better soft switching condition, Isw should lag the voltageVsw to a certain degree. For example, the wireless power transfer systemshould have just enough inductive reactive power at the output port ofthe power amplifier 1004 so that the voltage across S1 and S2 can bereduced to a level equal to or approximately equal to zero prior to theturn-on transitions of S1 and S2.

In accordance with the first principle, the efficiency improvement maybe achieved by monitoring the voltage across power switch S1 and/or thevoltage across power switch S2. More particularly, monitoring thevoltage across a power switch (e.g., S2) includes finding whether thevoltage across the power switch is reduced to a voltage equal to orapproximately equal to zero before a turn-on gate drive signal isapplied to the gate of the power switch.

According to the operating principles of a Class D power amplifier, thecontrol timing of the upper switch S1 and the lower switch S2 should besymmetrical. However, it may be desirable to monitor the voltage acrossone switch (e.g., the voltage across the lower switch S2) of the Class Dpower amplifier. To ensure a reliable and robust operation, the controltiming of S1 and S2 may be adjusted to be slightly asymmetrical so thatit is easy for S1 to achieve soft-switching. As such, it's not necessaryto monitor the voltage of S1. For example, the conduction period of S1in a switching cycle can be made slightly longer than that of S2. As aresult, S1 will achieve a zero voltage turn-on when S2 still experienceshard switching during a same operation mode. In other words, when S2achieves zero voltage switching, S1 achieves zero voltage switching toobecause the system configuration above ensures that it is easy for S1 toachieve zero voltage switching in comparison with S2.

It should be noted that monitoring the voltage across S2 to achieve zerovoltage switching for both S1 and S2 is merely an example. One skilledin the art will recognize that in alternative embodiments, achievingzero voltage switching for S1 and S2 can be accomplished by monitoringthe voltage across S1.

In accordance with the second principle of achieving soft switching, theefficiency improvement may be achieved by monitoring Isw shown in FIG.10. In some embodiments, the peak value or rms value of Isw can be usedas an indicator for achieving soft switching. Alternatively, the valueof Isw at the instants of turning on/off S1 and/or S2 can also be usedas a current measurement signal for the purpose of achieving better softswitching.

In some embodiments, the efficiency may be improved by turning on S1 andS2 when Isw is equal to or approximately equal to zero. To minimize theeffect of switching noise on monitoring the soft-switching condition ofS2, it may be better to measure the instantaneous value of Isw either atan instant right before the turn-off of S1 or at an instant right afterthe turn-on of S2. Furthermore, a suitable offset may be added into themeasured current signal so that a better insight into the current at theswitch turn-on instant and/or the switch turn-off instant may beobtained.

Depending on different designs and applications, it may be hard to getaccurate and clean current measurements at high frequencies due to thenoisy environment in or around a power amplifier. As an alternative,soft switching through monitoring Isw can be replaced by assessing thevoltage across S1 or S2 during the switching process. Assuming thevoltage across S2 (Vsw) is used for this purpose and the turn-on processof S2 is used as an example, there may be three scenarios to beconsidered. These three scenarios will be described in detail below.

In a first scenario, referring back to FIG. 7, Vsw is reduced to avoltage approximately equal to zero before S2 is turned on. As shown inFIG. 7, the derivative of Vsw is high right before S2 is turned on or inthe time period during which the body diode of S2 starts to conduct. Itshould be noted that the actual value of Vsw should be negative. Theabsolute value of Vsw is approximately equal to zero. Such a highderivative value of Vsw indicates that Isw is too high at the turn-on ofS2 or at the turn-off of S1. In other words, the inductive reactivepower is relatively high in the system. It should be noted that Iswshould be positive during this process to achieve soft switching for S2.FIG. 12 shows Isw is relatively high and the period of body diodeconduction (body diode of S2) is relatively long. Crt or Crr may beadjusted to reduce the inductive reactive power at the output port ofthe power amplifier. For example, the capacitance of Crt may be reducedin order to reduce the inductive reactive power at the output port ofthe power amplifier.

In a second scenario, Vsw is reduced to a voltage approximately equal tozero before S1 is turned on, and the derivative of Vsw is low orapproximately equal to zero right before S2 is turned on or the bodydiode of S2 starts to conduct. Such a low derivative of Vsw indicatesthat the current Isw is at a right value for achieving soft switching.It is not necessary to further adjust Crt and/or Crr. The waveformsassociated with the second scenario are illustrated in FIGS. 13-14.

In a third scenario, Vsw is at a significant value when S1 is turned on,and soft-switching is not achieved. This indicates that current Isw istoo small at the switching instants (could even be negative), and thereactive power at the output port of the power amplifier is either toosmall, or is capacitive. Crt or Crr should be adjusted to increase theinductive reactive power at the output port of the power amplifier. Forexample, the capacitance of Crt should be increased so as to increasethe inductive reactive power at the output port of the power amplifier.

It should be noted that the dead-time between S1's conduction and S2'sconduction may also be adjusted to achieve better soft-switchingresults. This can be performed in synch with the capacitance modulationtechnique described above if necessary.

As described above, the derivative of the voltage signal Vsw can be usedto indicate whether soft-switching has been achieved or adjustmentsbased upon the capacitance modulation technique are needed. Thederivative of the voltage signal Vsw can be obtained through a softwaremeans and/or a hardware means. The software means can be implemented asa digital differentiator and the like. The hardware means can beimplemented as a RC network, a combination of a RC network and anoperational amplifier, and the like. The software means and hardwaremeans described above are well known in the industry, and hence are notdiscussed in detail herein to avoid repetition.

In some embodiments, if the value of a voltage across a switch (e.g.,Vsw) and its derivative are both at a value equal to or approximatelyequal to zero at the instant when the switch is turned on, a bettersoft-switching condition has been achieved. As such, the bettersoft-switching condition can be determined by adding a soft-switchingobserver (not shown but illustrated in FIG. 11) to monitor the voltageacross the switch (e.g., Vsw) and/or the current flowing through theswitch (e.g., Isw). Based upon the capacitance modulation schemes above(three scenarios), the values of Crt and/or Crr may be adjustedaccordingly.

The output of the soft-switching observer may determine whether thesystem should increase or decrease the capacitances of Crt and/or Crr.Furthermore, the adjustment step depends on the output of thesoft-switching observer. For example, the step of the adjustment candepend on the value of Isw or the derivative of Vsw at the previousturn-on instant of S2. A filter may be included to filter out possiblenoise depending on the values of Isw and/or the derivatives of Vsw atthe last several turn-on instants of S2.

The output of the soft-switching observer may be constructed in thefollowing manner. First, the output of the soft-switching observergenerates zero when the power amplifier has already achieved softswitching and no further adjustment is necessary. Second, the output ofthe soft-switching observer generates a positive value when the poweramplifier has already achieved soft switching, but the inductivereactive power of the wireless power transfer system is too high (theswitch current is too high when the switch is turned on). The value ofthe positive value generated by the soft-switching observer indicatesthe adjustment speed and/or the adjustment step. Third, the output ofthe soft-switching observer generates a negative value when the poweramplifier has not achieved soft switching yet and the inductive reactivepower of the wireless power transfer system is too low and needs to beincreased. The absolute value of the negative value indicates theadjustment speed and/or the adjustment step.

In some embodiments, Isw can be sampled in synch with the switching ofthe power switches of the power amplifier. The sampled signal may befurther analyzed based upon the operating conditions when the samplingoccurs. For example, the sampled signal of Isw may be analyzed inconsideration with whether the switches are turned on withsoft-switching when the sampling occurs. Furthermore, filteringfunctions may be employed in the sampling process or the analyzingprocess, to further reduce noise and get an appropriate output.

It should be noted the output of the soft-switching observer may beconstructed in a variety of ways. For example, an offset may be added tothe output of the soft-switching observer so that the system avoidsdealing with a negative value at the output of the soft-switchingobserver. It should further be noted that the adjustment of Crt and/orCrr above can be accomplished either in a digital fashion or in ananalog fashion.

The capacitance modulation technique described above can be applied toother power amplifier topologies such as push-pull power amplifiers,class-E power amplifiers and the like. By directly assessing andregulating the soft-switching condition of a power switch in the poweramplifier through current information at a switch's turn-on instant orturn-off instant, a better soft-switching condition can be achievedagainst operating condition variations such as input voltage variations,output load variations, temperature variations, switch parasiticcapacitance variations and switching parameter variations, and circuitparameter variations including inductance and capacitance variations aswell as coupling variations.

The impact of the EMI filters (e.g., transmitter EMI filter and receiverEMI filter) and other auxiliary circuits (e.g., impedance matchingcircuit) on the switches' soft-switching may also be considered when thesoft-switching observer generates its output indicating the capacitancemodulation step and speed. In sum, through the capacitance modulationtechnique described above, the system not only achieves soft-switchingof the power switches (and thus lower power losses, higher devicereliability, and lower noise), but also ensure the reactive power andcurrent stresses are at the minimum while delivering the required outputpower. One advantageous feature of having the capacitance modulationtechnique described above is that high performance and low cost can beachieved in a design at the same time.

In the system shown in FIG. 10, there may be a feedback control systemhaving two control variables, namely the capacitance of Crt and thecapacitance of Crr. Two outputs of the control system may be controlledthrough adjusting these two control variables. In some embodiment, oneoutput of the control system is the output voltage or the output powerof the receiver. The other output of the control system is thesoft-switching condition of the power switches.

In some embodiments, feedback control mechanisms can be used todetermine the values of the control variables. According to the feedbackcontrol mechanisms, the control system outputs can be used as inputs ofa feedback controller. The modulation of Crr changes both the real partand the imaginary part of the impedance of the receiver power circuitreflected in the transmitter, and thus affects both active power andreactive power in the wireless power transfer system. The modulation ofCrt only changes the imaginary part of the impedance in the transmitterpower circuit. In addition, the modulation of Crt changes the magnitudeof the current flowing through the transmitter coil, thereby affectingboth reactive power and active power in the wireless power transfersystem. The changes of the capacitance of Crt and/or the capacitance ofCrr will cause both active power changes, which contribute to thechanges of Vo or Po, and reactive power changes, which contribute to thechange of the soft-switching condition. Therefore, two outputs in thisfeedback control system exist. In some embodiments, a two-input andtwo-output controller may be employed to fulfill the feedback functionsdescribed above. Such a two-input and two-output controller may be builtwith information transferred across the Bluetooth communication channel.In alternative embodiments, a fast speed control mechanism may beimplemented by using a plurality of local controller in the transmitterand/or in the receiver (receivers).

FIG. 11 illustrates a block diagram of an implementation of a feedbackcontrol system of a wireless power transfer system in accordance withvarious embodiments of the present disclosure. The wireless powertransfer system 1100 comprises a power transmitter 1102 and a powerreceiver 1112 coupled together through magnetic coupling. It should benoted while FIG. 11 shows one power receiver is coupled to the powertransmitter 1102, multiple power receivers may be alternatively includedin the wireless power transfer system 1100.

The power transmitter 1102 comprises a soft switching observer 1104, acontrol and protection unit 1106 and a transmitter Bluetoothcommunication unit 1108. As shown in FIG. 11, the soft switchingobserver 1104 may receive two input signals, namely Isw and Vsw andgenerates an output coupled to a second input of the control andprotection unit 1106. The soft switching observer 1104 may also use thegate timing information of S1 and S2, which is generated internally inthe transmitter control system. The control and protection unit 1106have a first input receiving Vin, a third input receiving the current Itflowing through the transmitter coil and a fourth input receiving avoltage Vcrt across Crt. The control and protection unit 1106 has aninput/output connected to the transmitter Bluetooth communication unit1108. The control and protection unit 1106 has a first output forcontrolling the gate timing of S1 and S2, and a second output formodulating the capacitance of Crt.

The power receiver 1112 comprises a voltage/power regulator 1114, aprotection unit 1116 and a receiver Bluetooth communication unit 1118.As shown in FIG. 11, the voltage/power regulator 1114 may receive twoinput signals, namely Vo and Io. The voltage/power regulator 1114 mayalso receive a signal from the receiver Bluetooth communication unit1118 and/or the protection unit 1116. The voltage/power regulator 1114generates an output signal fed into a first input of the protection unit1116. The protection unit 1116 has a second input receiving the currentIr flowing through the receiver coil and a third input receiving avoltage Vcrr across Crr. The protection unit 1116 has an input/outputconnected to the receiver Bluetooth communication unit 1118. Theprotection unit 1116 may have a first output for controlling the gatetiming of D1 and D2 when the rectifier is implemented as a synchronousrectifier, and a second output for modulating the capacitance of Crr. Itshould be noted that the gate timing of D1 and D2 is applicable to D1and D1 only when the rectifier formed by D1 and D2 is replaced bysynchronous rectifier. The receiver Bluetooth communication unit 1118may communicate to the transmitter Bluetooth communication unit 1108 asshown in FIG. 11.

As shown in FIG. 11, the soft switching observer 1104 receives thedetected signals Isw and/or Vsw. The soft switching observer 1104decides whether an adjustment is needed to achieve a soft switchingcondition based on the information received at the inputs (e.g., Vswand/or Isw). The control and protection unit 1106 may comprise acontroller. The controller receives the output signal from the softswitching observer 1104 as well as the input voltage Vin, the currentflowing through the transmitter coil Lt and the voltage across Crt.Based upon the received signals, the controller in the control andprotection unit 1106 may adjust the capacitance of Crt and/or the timingof S1 and S2 at the same time when necessary. The adjustments of thecapacitance of Crt and/or the timing of S1 and S2 help to archive thebetter soft switching of S1 and S2.

The controller in the control and protection unit 1106 may comprise afeedback compensator such as a proportional-integral-derivative (PID)compensator. The PID compensator is configured such that the output ofthe soft switching observer 1104 generates a value equal to zero or afixed value representing a soft switching condition.

The controller in the control and protection unit 1106 may beimplemented as a digital controller. The digital controller may beimplemented in hardware, software, any combinations thereof and thelike. For example, the controller may be implemented as an adder withsome filtering functions, a lookup table with some filter functions totranslate the output of the soft-switching observer 1104 into acapacitance value (or states of the controllable switches in a variablecapacitance network). The controller may also take into account Vin andadjust the capacitance value of Crt accordingly in a feedforward manner.

In some embodiments, Crt can also be used to protect the powertransmitter 1102 from a variety of abnormal operating conditions such asover-voltage, over-current, over-temperature and any other faultyconditions. For example, the system may keep monitoring the currentflowing through the transmitter coil and/or the voltage across theresonant capacitor Crt. When an over current or an over voltage occurs,the system may be protected by changing the capacitance value of Crt.Depending on design needs and different applications, the value of Crtcan be adjusted to a small or a big value so that the power and currentwill be reduced quickly.

The transmitter Bluetooth communication unit 1108 can pass informationbetween the power transmitter 1102 and the power receiver 1112.Furthermore, the communication between the power transmitter 1102 andthe power receiver 1112 helps to adjust the control parameters andfunctions slowly in the power transmitter 1102 and the power receiver1112.

In the power receiver 1112, a voltage/power regulator 1114 regulates theoutput voltage or power to a desired value based on the detected signalsVo and/or Io. This regulation can be performed by a feedbackcompensator, such as a PID compensator, inside the voltage/powerregulator 1114. Furthermore, a feedforward control mechanism forcontrolling Io and/or Vo may be also used at the same time.

It should be noted that the control mechanism described above is merelyan example. There may be many alternatives, modifications and variationsfor implementing the control schemes for the voltage/power regulator1114. For example, this regulation can be done by a search mechanism tosearch an appropriate value for the capacitance of Crr. Such anappropriate value helps the power receiver 1112 achieve better results.

It should further be noted that there may be more than one capacitanceof Crr obtained during the search mechanism. In other words, multiplevalues of Crr may give similar results (e.g., output power or outputvoltage) in a complex system. Among these values of Crr, the appropriateone is the capacitance having a resonant frequency close to the systemfrequency. Such a resonant frequency is alternatively referred to as thereceiver resonant point throughout the description.

In some embodiments, the search mechanism may be implemented by startingfrom a value at or close to the receiver resonant point in any searchaction of Crr and performing new searches regularly to avoid enteringdeeply into a wrong search direction. Similarly, the initial output ofthe feedback compensator in the voltage/power regulator 1114 can be setto a value corresponding to a suitable Crr value. The suitable Crr valuemay result in a resonant frequency close to the receiver resonant point.Furthermore, the regulator can be reset to the initial value regularly.

In some embodiments, Io can be sensed from the rectifier, at the outputport of the power receiver and/or from the load. In some embodiments,there may be a communication channel between the load and the powerreceiver. The control mechanism of the power receiver may be coordinatedwith any changes in the load through the communication channel betweenthe load and the power receiver.

The output of the voltage/power regulator 1114 is used to modulate thecapacitance value of Crr. In some embodiments, the modulation of thecapacitance of Crr can also be used to protect the receiver from avariety of abnormal operating conditions such as over-voltage,over-current, over-temperature and other abnormalities. For example, thecurrent flowing through the receiver coil, the voltage across theresonant capacitor Crr, the output voltage Vo and/or the output currentIo may be monitored by the power receiver 1112. When a fault (e.g., overcurrent or an over voltage) occurs, the system may be protected bychanging the capacitance value of Crr. Depending on design needs anddifferent applications, the capacitance value of Crr can be adjusted toa small or a big value so that the power and current will be reducedquickly.

The receiver Bluetooth communication unit 1118 can pass informationbetween the power transmitter 1102 and the power receiver 1112.Furthermore, the communication between the power transmitter 1102 andthe power receiver 1112 helps to adjust the control parameters andfunctions slowly in the power transmitter 1102 and the power receiver1112.

To avoid severe interaction between the transmitter control and thereceiver control, the local feedback control loops in the transmitterand in the receiver should have different control speeds. For example,the transmitter can modulate Crt with a first control bandwidth. Thereceiver can modulate Crr with a second control bandwidth. In order toavoid interference between these two control loops, the control systemshould be designed such that the first control bandwidth is higher thanthe second control bandwidth.

The coordination of the Crt capacitance modulation and the Crrcapacitance modulation can be achieved without passing significantinformation through the slow Bluetooth communication channel. However,if necessary, adjustment and calibration information can pass throughthe Bluetooth communication channel to further improve the performanceof the local control loops.

FIG. 12 shows a variety of waveforms associated with a wireless powertransfer system with weak coupling between the transmitter and thereceiver coils in accordance with various embodiments of the presentdisclosure. In some embodiments, the coupling coefficient is about 10%.

The horizontal axis of FIG. 12 represents intervals of time. The unit ofthe horizontal axis is microsecond. There may be four vertical axes. Thefirst vertical axis Y1 represents the voltage (V_(SW)) across thedrain-to-source of the switch S2, the gate drive voltage (V_(s2g)) ofthe switch S2 and the output voltage Vo. The second vertical axis Y2represents the current (I_(SW)) flowing through the first inductor L1.The third vertical axis Y3 represents the current (I_(t)) flowingthrough the transmitter coil Lt and the voltage (V_(crt)) across thecapacitor Crt. The fourth vertical axis Y4 represents the current(I_(r)) flowing through the receiver coil Lr and the voltage (V_(crr))across the capacitor Crr.

As shown in FIG. 12, the output voltage Vo is maintained around 5 V inorder to be compatible with the USB specification. From t1 to t2, Vsw isapproximately equal to −1V. The negative voltage of Vsw indicates theconduction of the body diode of S2. From t3 to t4, the Vsw is about onediode voltage drop above the voltage rail (e.g., Vsw in steady state).The one diode voltage drop indicates the conduction of the body diode ofS1. Since both body diodes conduct before turning on their respectiveswitches, both S1 and S2 achieve soft switching. However, the conductiontime of the body diodes is too long. As a result, the efficiency of thewireless power transfer system may be not as good as what it is expectedto be under a soft switching operation condition.

FIG. 13 shows a variety of waveforms associated with a wireless powertransfer system with the same coupling as that of FIG. 12 in accordancewith various embodiments of the present disclosure. In some embodiments,the output power of the wireless power transfer system is about 33 W.The coupling coefficient is about 10%. In comparison with the systemshown in FIG. 12, the capacitance values of Crt and Crr are adjusted toachieve both output voltage regulation (Vo is slightly greater than 5 V)and better soft switching of S1 and S2.

As shown in FIG. 13, the waveform of Vsw indicates that both S1 and S2are turned on with soft switching. The turn-on time of S1 is not equalto the turn-on time of S2. Under this asymmetric operation of S1 and S2,it is easier for S1 to enter into soft switching in comparison with S2.The waveform of Vsw from t1 to t2 indicates the body diode of S1conducts for a little while before S1 is turned on. In comparison withthe body diode conduction of S1 in FIG. 11, the conduction time shown inFIG. 13 is shorter. As a result, a better soft switching condition hasbeen achieved. It should be noted that during the turn-on transitions ofS1 and S2, the current Isw is much lower than its peak value as shown inFIG. 13. Such a low current helps to reduce the switching losses of S1and S2.

FIG. 14 shows a variety of waveforms associated with a wireless powertransfer system with stronger coupling between the transmitter and thereceiver coils in accordance with various embodiments of the presentdisclosure. In some embodiments, the output power of the wireless powertransfer system is about 33 W. The coupling coefficient is about 25%.

In comparison with the system shown in FIG. 12, the capacitance valuesof Crt and Crr are adjusted to achieve both output voltage regulation(Vo is slightly greater than 5 V) and better soft switching of S1 andS2. As shown in FIG. 14, the waveform of Vsw indicates that both S1 andS2 are turned on with soft switching. The turn-on time of S1 isapproximately equal to the turn-on time of S2. In other words, the poweramplifier is under a symmetric operation. The waveform of Vsw indicatesboth the body diode of S1 and the body diode of S2 barely conduct. As aresult, a better soft switching condition has been achieved. It shouldbe noted that during the turn-on transitions of S1 and S2, the currentIsw is approximately equal to zero as shown in FIG. 14. Such a lowswitching current helps to reduce the switching losses of S1 and S2.

One advantageous feature of having the capacitance modulation techniqueis both better soft switching and tight output regulation may beachieved by adjusting the capacitances of Crt and Crr. Moreparticularly, the adjustment of Crr is used to regulate the outputvoltage or power. The adjustment of Crt is used to maintain a bettersoft switching condition as shown in FIGS. 13-14. In sum, thecapacitance modulation technique helps to improve the system efficiencyand reduce the system cost.

In some embodiments, the modulation of Crt and/or Crr may be applicableto a soft-start process of a wireless power transfer system. Forexample, the initial value of Crr can be set to a value that resultingin very low or zero output power. The initial value of Crr is usuallyaway from the value that generates the receiver resonant point when Crrresonates with Lr.

The capacitance value of Crr gradually changes towards the capacitancevalue generating the receiver resonant point. At the same time, theoutput power of the rectifier also gradually increases. The gradualincrease of the output power fulfills the soft-start process of thewireless power transfer system. In some embodiments, Crr can stop at anappropriate value. The selection of this appropriate value is incoordination with the output power and/or voltage regulation.

The gradual change of Crr can also be used to identify the actualresonant point of the receiver. For example, when the receiver operatesat the receiver resonant point, the output power reaches its highestlevel for a given transmitter current (e.g., the ratio of Po to the rmsvalue of It is maximized). This ratio can also be used to find theactual mutual inductance between the transmitter coil and the receivercoil because there is a well-defined relationship between the mutualinductance and the reflected resistance.

In some embodiments, the mutual inductance information may be used toimprove the system performance. The receiver resonant point informationcan be used to limit the range of capacitance modulation. It isdesirable to modulate Crr only in one side of the receiver resonantpoint in normal operation. For example, it is desirable to only allowCrr to be modulated to a value less than the capacitance generating thereceiver resonant point. The mutual inductance between the receiver andthe transmitter in a wireless power transfer system may change when therelative position of the transmitter coil and the receiver coil changes.Furthermore, other factors such as metal or magnetic objects placednearby may change their relative position between the receiver and thetransmitter. In order to solve the issue caused by the change of therelative position, the receiver resonant point and the mutual inductancecan be retested regularly or when any related relative position changeoccurs around the receiver.

The mutual inductance is sensitive to foreign objects, so the test ofthe mutual inductance can be a good way to identify the presence of aforeign object close to the receiver or the transmitter. Similarly, themethod of testing the mutual inductance described above can be used totest the transmitter resonant point. With the resonant capacitance ofevery receiver coupled to the transmitter reduced to a levelapproximately equal to zero, the receiver coils may not generateconsiderable currents to interact with the transmitter's magnetic field.However, the change of the transmitter coil's inductance caused bymagnetic parts and/or metal parts placed around the receivers stillexists. By sweeping the value of Crt slowly and measuring thetransmitter current or an impedance associated with the resonant tank ofthe transmitter, the transmitter resonant point, at which Crt and Ltresonate at the system frequency, can be identified.

During some phases of operation of a wireless power transfer system, atransmitter needs to identify the presence of a valid receiver withouttransferring significant power to the receiver. During this kind ofsystems, Crr can be set very low or even zero, so the power delivered tothe receiver's output is very low, but the voltage across Lr can be highenough so that a different power path (also coupled to Lr) can deliverenough energy to wake up the receiver's controller and enable acommunication between the receiver and the transmitter.

FIG. 15 illustrates a schematic diagram of a variable capacitancenetwork in accordance with various embodiments of the presentdisclosure. The variable capacitance network 1500 is used to adjust thecapacitance of Crr. In other words, Crr shown in FIG. 10 may be replacedby the variable capacitance network 1500 shown in FIG. 15. It should benoted that, in order to achieve capacitance modulation, Crt shown inFIG. 10 may be replaced by a variable capacitance network similar tothat shown in FIG. 15.

The variable capacitance network 1500 comprises a diode Dx, a dividerformed by Rx1 and Rx1, a first capacitor Cr0 connected in parallel withRx2 and a plurality of capacitor-switch networks. As shown in FIG. 15,there may be five capacitor-switch networks. A first capacitor-switchnetwork comprises a capacitor Cx0 and a switch Sx0 connected in seriesand further connected in parallel with Rx1 as shown in FIG. 15.Likewise, a second capacitor-switch network comprises a capacitor Cx1and a switch Sx1 connected in series and further connected in parallelwith Rx1; a third capacitor-switch network comprises a capacitor Cx2 anda switch Sx2 connected in series and further connected in parallel withRx1; a fourth capacitor-switch network comprises a capacitor Cx3 and aswitch Sx3 connected in series and further connected in parallel withRx1; a fifth capacitor-switch network comprises a capacitor Cx4 and aswitch Sx4 connected in series and further connected in parallel withRx1.

It should be recognized that while FIG. 15 illustrates the variablecapacitance network 1500 with five capacitor-switch networks, thevariable capacitance network 1500 could accommodate any number ofcapacitor-switch networks. It should further be noted that the firstcapacitor-switch network may be replaced by a capacitor depending ondifferent design needs and applications.

The diode Dx functions as a clamp diode. In some embodiments, the diodeDx may be replaced by a unidirectional switch having an anti-parallelbody diode such as a MOSFET. The switches Sx0, Sx1, Sx2, Sx3 and Sx4 areused to control the total capacitance of the variable capacitancenetwork 1500. By controlling the on/off states of the switches Sx0, Sx1,Sx2, Sx3 and Sx4, a variety of capacitances may be obtained accordingly.For example, when all switches Sxo through Sx4 are open, the equivalentcapacitance of the variable capacitance network 1500 is approximatelyequal to zero. On the other hand, when all switches Sxo through Sx4 areclosed, the equivalent capacitance of the variable capacitance network1500 reaches its maximum capacitance value.

In some embodiments, the diode Dx may be implemented as a switch. Whenthis switch is closed, the equivalent capacitance of this variablecapacitance network may be approximately equal to the capacitance ofCr0.

Rx1 and Rx2 are used to ensure the current flowing through Dx is smallduring normal operation. The value of the current flowing through Dx maybe adjusted by selecting the values of Rx1 and Rx2. In some embodiments,Rx1 and Rx2 may be two separate components. In alternative embodiments,Rx1 and Rx2 may be from the parallel parasitic resistance of thecorresponding capacitors (e.g., Cr0 and Cx0) and switches (e.g. Sx0).

As described above, the capacitance of the variable capacitance network1500 may be reduced to a level approximately equal to zero by keepingall switches open. Such a feature may help wireless power transfersystems (e.g., the wireless power transfer system shown in FIG. 10)achieve flexible protection and/or standby control mechanisms. Moreparticularly, in wireless power transfer systems (e.g., the wirelesspower transfer system shown in FIG. 10), by reducing the capacitance ofthe variable capacitance network 1500, the output power of the receivermay drop accordingly no matter how much current is flowing through thetransmitter coil.

In a standby mode of a receiver (e.g., the receiver shown in FIG. 10),the capacitance of Crr is set to a minimum value (approximately equal tozero) by controlling the variable capacitance network 1500. Since thecapacitance of Crr is very small, the currents in the receiver coil, thereceiver resonant circuit and the rectifier are reduced dramatically. Asa result, almost no power is delivered from the receiver to the loadcoupled to the receiver. If only one receiver is magnetically coupled tothe transmitter and the one receiver is set to operate in the standbymode described above, the current flowing through the transmitter isvery small because the one receiver operates in the standby mode. Underthis standby mode, the reactive power in the transmitter is able tomaintain a better soft switching condition for S1 and S2.

In some embodiments, the efficiency of a wireless power transfer systemmay be further improved by applying an active mode and a standby mode inan alternating manner to the wireless power transfer system at afrequency much lower than the system frequency. This is similar to aburse-mode operation in a conventional PWM power supply. During theactive mode and the standby mode, both the receiver resonant capacitanceand the transmitter resonant capacitance are set to suitable values(e.g., capacitance values having a resonant frequency close to theresonant point). As such, some power is transferred from the transmitterto the receiver. During a standby mode, the capacitance of the receiverresonant capacitor or the transmitter resonant capacitor is adjusted toa value far away from their respective resonant points. As a result,there is little power transferred between the transmitter and thereceiver. Furthermore, the output power and/or voltage may be regulatedby controlling the duty cycle during the active mode. It should be notedthat some other suitable efficiency improvement methods such asmeasurement, calibration, detection and the like may be performed duringthe standby mode operation. As a result, the system performance can befurther improved.

The standby mode control mechanism described above may be applicable toa system having multiple receivers. In some embodiments, multiplereceivers are magnetically coupled to a transmitter and the sum of thetotal power demand from all receivers exceeds the power capability ofthe transmitter, the standby mode operation described above can be usedto regulate and limit the power delivered to each receiver or somereceivers. As a result, the transmitter can work within its safeoperating region while delivering power to receivers in an acceptablemanner.

Alternatively, the transmitter may instruct some or all receivers tomodify their power demands through the Bluetooth communication systemshown in FIG. 10. In response to the instruction from the transmitter,the related receivers can reduce their power demands by modulating theirresonant capacitances Crr. Furthermore, because the modulation of thecapacitance of Crr in a receiver changes the reflected impedance of thereceiver in the transmitter, the power distribution among the receiverscan be adjusted by modulating the capacitance of Crr in the receivers.

In sum, the output power and/or voltage of each receiver can becontrolled separately in a smooth manner by employing the capacitancemodulation technique in the receivers. The ability of reducing the powerin any receiver to a level approximately equal to zero without affectingthe operation of other receivers gives flexibility to achieve bettersystem operation and protection control mechanisms. In particular, themodulation of the capacitance of Crr in a receiver may be used toachieve a better soft-start process and/or a better soft-stop process ofthe receiver as discussed earlier. As such, the addition or removal of areceiver magnetically coupled to the transmitter may have minimumoperational impacts on other receivers magnetically coupled to thetransmitter by employing the capacitance modulation technique.

In many applications, the input voltage Vin of a power amplifier (e.g.,power amplifier shown in FIG. 10) may not have a fixed voltage. Forexample, if Vin is from an output of an ac/dc power supply or a dc/dcpower supply, Vin may be advantageously changed by coordinating theoperation of the ac/dc power supply or the dc/dc power supply with thewireless power transfer system (e.g., wireless power transfer systemshown in FIG. 10).

In some embodiments, the ac/dc power supply or the dc/dc power supplymay be part of the wireless power transfer system to facilitate thecoordination process described above. In particular, during thecoordination process, Vin can be used to control the output power and/orthe output voltage of a receiver of the wireless power transfer system.As such, there may be one more control variable. This additional controlvariable can be used to further improve the performance of the wirelesspower transfer system. For example, after having the additional controlvariable, the modulation of the capacitance of Crr in a receiver can beused to fine-tune the resonant frequency of the receiver resonant tankto a frequency approximately equal to the system frequency. As a result,the current in the transmitter coil can be reduced.

In some embodiments, multiple receivers may be magnetically coupled tothe transmitter. The fine-tune process of the resonant frequency of areceiver described above can be applied to the one with a maximum powerdemand or the one requiring the highest transmitter current.

Furthermore, after having the additional control variable Vin, in awireless power transfer system having a single receiver, the capacitancevalue of Crr can be fixed to a suitable value (e.g., a value having aresonant frequency around the receiver resonant point) to maintain goodperformance. Such a fixed capacitance of Crr helps to simplify thereceiver design, thereby reducing the cost of the wireless powertransfer system. It should be noted that after having the additionalcontrol variable, it is still necessary to modulate the capacitance ofCrt in the transmitter to maintain a better soft-switching condition forpower switches S1 and S2. The presence of the additional controlvariable Vin may help to reduce the variation range of the capacitanceof Crt.

In some embodiments, the additional control variable Vin can also beused to limit the current in the transmitter coil Lt. Because thecurrent flowing through the transmitter coil Lt plays an important rolein determining the coil temperature, the power losses, the system EMIperformance, the magnetic field strength around the transmitter coil andthe like, it is desirable to limit the transmitter coil current to alower value. In some embodiments, when the wireless power transfersystem demands more power, the additional control variable Vin can beused to increase the total power available for a given set of circuitparameters. On the other hand, when the wireless power transfer system'spower demand drops, Vin can be reduced accordingly to achieve bettersystem efficiency. In this way, the additional control variable Vinhelps to maintain a better operating condition over a wide range ofoutput power and a variety of coupling conditions. Furthermore, when awireless power transfer system needs to be shut down for protectionpurposes, Vin can be reduced to a very low value or zero to shut downthe wireless power transfer system in a smooth manner.

In some embodiments, the output voltage Vo may be used as one additionalcontrol variable for further improving the performance of a wirelesspower transfer system. For example, if the output voltage Vo is not useddirectly to power a sensitive load (e.g., power is delivered to abattery through a battery charger or power is delivered to a loadthrough one or more power converters), Vo can be adjusted within acertain range to achieve a better system operation condition. Moreparticularly, the control variable Vo may be used to better compensatethe coupling coefficient variations. When the coupling between areceiver and a transmitter is strong, the output voltage Vo is set to ahigher value in response to the strong coupling between the transmitterand the receiver. On the other hand, when the coupling between thereceiver and the transmitter is weak, the output voltage Vo is set to alower value in response to the weak coupling between the transmitter andthe receiver. The control mechanism based upon the Vo adjustment helpsto reduce the maximum current in the transmitter coil as well as thecapacitance ranges of Crt and Crr.

In some embodiments, a variable output voltage Vo in a certain range maybe acceptable. In order to limit the stresses on the power components,Vo may vary in response to output power changes. Furthermore, it may bedesirable to set a higher Vo for a receiver when the power required fromthe receiver is high. As such, the currents in the receiver coil andother components as well as the current in the load are limited to areasonable value to achieve better performance, thereby reducing thecost of the system.

In some embodiments, when the power demand from a receiver is very low,it may be desirable to operate the transmitter and/or the receiver in aburst mode operation. The burst mode can be achieved by combining anactive mode with a standby mode. For example, when the required powerfrom a receiver is below a certain threshold, the system is in a normalactive power transfer mode for a certain time, and then enters into astandby mode for another period of time. The standby mode can be createdthrough a variety of methods including reducing the input voltage fedinto the power amplifier to a very low voltage level, changing thecapacitance of Crt to a much lower value or much higher value, and/orchanging the capacitance of Crt to a much lower value or much highervalue. It should be noted that either a higher value of the resonantcomponents or a lower value of the resonant components may help thesystem achieve the standby mode because the output power is low when theresonant frequency is away from the system frequency.

The burst mode described above is also applicable to a wireless powertransfer system having multiple receivers magnetically coupled to atransmitter. In particular, any receiver can enter into a burst modeoperation by modulating the capacitance of Crr of this receiver withoutinterrupting the operation of other receivers. Furthermore, thetransmitter can be put into the burst mode operation by modulating thecapacitance of Crt if all of the receivers magnetically coupled to thetransmitter are required to generate low power and operate in the burstmode. Under such a burst mode operation, the output power of eachreceiver relative to other receivers can still be regulated by selectinga right value for the resonant capacitor in the receiver or by adjustingthe duty cycle of the active power transfer mode of the receiver.

In a wireless power transfer system, the voltages and currents in thetransmitter and the receiver are all affected by the changes of theresonant capacitances in the transmitter and in the receiver. Thecapacitance modulation in the transmitter and the receiver can be usedto provide an in-band communication channel between the transmitter andthe receiver. In order to achieve this in-band communication channel, anappropriate communication protocol needs to be established to facilitatethe in-band communication in a wireless power transfer system. Thecurrent, voltage and/or power in the transmitter and the receiver of thewireless power transfer system can be used as the means to conveyinformation for the in-band communication.

It should be noted that if the operating frequency of a wireless powertransfer system is allowed to change, the operating frequency of awireless power transfer system can be used as a control variable. Theoperating frequency variations can be used to control and protect thewireless power transfer system in a way similar to the control mechanismbased upon the Vin variations described above.

The control mechanisms described above may be applicable to a soft startprocess in a wireless power transfer system. For example, during a softstart process, the wireless power transfer system may operate in ahybrid mode comprising both an active mode and a standby mode. Inparticular, the active mode and the standby mode are applied to thewireless power transfer system in an alternating manner. In addition,the duty cycle of the active mode increases gradually during the softstart process. In this way, the average output power is controlled bythe duty cycle of the active mode operation.

During the soft start process, the capacitance modulation technique canbe used to control the output power of the wireless power transfersystem during the active mode operation. For example, by modulating thecapacitance, the output power in the active mode is set to a lower levelduring an early stage of the soft start process, and the output powerincreases gradually as the soft start process progresses towards thecompletion. In this way, the start-up process can be made even softerthan the process having only the capacitance modulation or the dutycycle control. It should be noted that other control variables such asgradually increasing the input voltage, gradually changing the switchingfrequency and the like can also be used to achieve a smooth soft startprocess. The other control variables can be taken either individually orin combination with the capacitance modulation and/or the duty cyclecontrol. Furthermore, it should be noted that all control methods for asoft-start process described above can be applicable to a soft-stopprocess if required.

To reduce the system cost, different parts of a wireless power transfersystem can be integrated into a plurality of ICs. The level ofintegration should be determined in consideration with the power leveland/or the system design requirements. For some application, the controlsystem of a transmitter and/or the control system of a receiver can beintegrated into one IC, which may also include the capacitancemodulation circuit and some of the variable capacitance networks shownin FIG. 15. For some applications, the power amplifier, the receiverrectifier, the EMI filter, and the capacitance modulation circuit, andsome of the variable capacitance networks can be integrated into theirrespective ICs separately.

In some embodiments, depending upon different applications and designneeds, a plurality of ICs comprising the power amplifier, the receiverrectifier, the EMI filter, the capacitance modulation circuit and someof the variable capacitance networks can be integrated into one ICthrough suitable semiconductor fabrication processes such as verticallystacking a plurality of ICs on top of each other and the like.

In some embodiments, depending upon different applications and designneeds, the whole transmitter except the transmitter coil may beintegrated into one IC, and the whole receiver except the receiver coilmay be integrated into another IC.

In some embodiments, depending different applications and design needs,the whole transmitter including the transmitter coil can be integratedinto one IC. The whole receiver including the receiver coil can beintegrated into another IC.

In sum, modulating reactive components in a transmitter (e.g., thecapacitance of Crt) and in a receiver (e.g., the capacitance of Crr) ofa wireless power transfer system has been described above. Thecapacitance modulation technique helps to improve the performance of thewireless power transfer system through modulating the resonanceprocesses in the transmitter and the receiver.

In some wireless power transfer systems, one or more additionalintermediate resonators, which may be a coil coupled to a resonantcapacitor, are placed between the transmitter coil and the receivercoil. The resonance modulation technique described above can also beapplied to the one or more intermediate resonators to achieve betterresults in a manner similar to that used for modulating the resonantcomponent in a transmitter (e.g., modulating the capacitance of Crt) orin a receiver (e.g., modulating the capacitance of Crr).

Modulating a capacitance or inductance to regulate the power processingcan be used in other configurations. For example, if the transmitter isimplemented as a current source, the capacitance of the resonantcapacitor of a receiver coupled to the transmitter can be modulated toregulate the receiver's output.

In a power supply having multiple outputs, the capacitance modulationtechnique described above can be used to regulate some outputs too. Forexample, the power supply may have a structure similar to that shown inFIG. 10 except that a plurality of receivers are strongly coupled to thetransmitter. In other words, the transmitter may be a primary side ofthe power supply and the plurality of receivers may form the secondaryside of the power supply. In operation, the capacitance in the primaryside and/or the switching frequency can be used to maintain a bettersoft-switching condition for the primary switches or to regulate one ofthe multiple outputs. The capacitance modulation in the secondary sidecan be used to regulate the other outputs.

In some embodiments, an input of a power converter is connected to an acpower supply (e.g., a 110 V ac voltage). The power converter convertsthe 110 V ac voltage from a wall socket into a dc voltage suitable tothe power amplifier of a wireless power transfer system. The powerconverter may be implemented as an ac/dc power adapter comprising anac/dc rectifier and a dc/dc converter. Alternatively, the powerconverter may be implemented as a dc/dc converter if the source power isa dc power supply. The techniques discussed above can be used to designthe dc/dc converter. For some low power applications, the topologiesdescribed below with respect to FIG. 16 may be alternatively used.

FIG. 16 illustrates a schematic diagram of a zero-voltage switchingasymmetric half-bridge converter in accordance with various embodimentsof the present disclosure. The zero-voltage switching asymmetrichalf-bridge converter 1600 comprises a primary side circuit comprisingS1, S2, C1 and C2, a secondary side circuit comprising D1, D2 and Co, afirst transformer T1 coupled between the primary side and the secondaryside and a second transformer T2 coupled between the primary side andthe secondary side. The zero-voltage switching asymmetric half-bridgeconverter 1600 further comprises a controller 1602 having a firstinput/output terminal generating a gate drive signal for S1, a secondinput/output terminal generating a gate drive signal for S2, a thirdinput/output terminal receiving a detected current signal Ip flowingthrough the primary side of the transformer winding of T1, a fourthinput/output terminal receiving a detected voltage signal Vsw across thedrain-to-source of S1, a fifth input/output terminal coupled to awinding of the first transformer T1 and a sixth input/output terminalcoupled to a winding of the second transformer T2.

In some embodiments, S1 and S2 are controlled by two complementary gatedrive signals. For example, S1 has a conduction duty cycle of D, and S2has a duty cycle of 1-D. It should be noted that there may be a shortdead-time between S1's conduction period and S2's conduction period ineach switching cycle.

The first transformer T1 comprises a primary winding T1 p, a secondarywinding T1 s and an auxiliary winding T1 a. The second transformer T2comprises a primary winding T2 p, a secondary winding T2 s, and anauxiliary winding T2 a. As shown in FIG. 16, T2 p and T1 p are connectedin series between the common node of S1 and S2, and the common node ofC1 and C2. T1 s is connected in series with D1; T2 s is connected inseries with D2. T1 a is coupled to the fifth input/output of thecontroller 1602 and T2 a is coupled to the sixth input/output of thecontroller 1602.

D1 and D2 form a rectifier coupled between T1 s and T2 s, and theoutput. D1 and D2 can be implemented as a synchronous rectifier. C1, C2,S1 and S2 form a half-bridge configuration. Co is an output capacitor tofurther attenuate the ripple of the output voltage Vo.

When S1 conducts, a positive voltage is applied to T2 p, and D2 conductsbecause it is forward-biased. The power is delivered to the outputthrough T2 s. During this time, a negative voltage is applied to T1 p.As a result, D1 is reverse-biased. The reverse-biased D1 prevents T1 sfrom delivering power to the output. The first transformer T1 acts likean inductor. In particular, the primary winding of the first transformerT1 has been charged and energy is stored in the first transformer T1.After a conduction period of DTs (Ts is the duration of a switchingcycle and D is the duty cycle of S1), S1 is turned off. A negativecurrent Ip charges the capacitance across S1 and discharges thecapacitance across S2. As a result, Vsw moves towards the positive rail.The voltage across S2 is approximately equal to zero because Vsw isapproximately equal to the positive rail. As such, S2 can be turned onwith zero voltage switching after a short transition time.

When S2 is turned on, a positive voltage is applied to T1 p. In responseto the positive voltage applied to T1 p, D1 starts to conduct and energyis delivered from T1 to the output. During this time, D2 isreverse-biased. The reverse-biased D2 prevents T2 s from deliveringpower to the output. The second transformer T2 acts like an inductor. Inparticular, the primary winding of the second transformer T2 has beencharged and energy is stored in the first transformer T2. The storedenergy will be delivered to the output in the next switching cycle.

After a conduction period of slightly less than (1-D)Ts, S2 is turnedoff. Ip, which is positive now, discharges the capacitance across S1.After the capacitor across S1 has been discharged, Vsw moves towards thenegative rail. After a short transition time, S1 can be turned on withzero voltage switching. In this way, both S1 and S2 can be turned onwith zero voltage switching, which helps to achieve a high efficiencyand low EMI operation of the zero-voltage switching asymmetrichalf-bridge converter 1600.

The turns ratio and magnetizing inductance of T1 and T2 can be designedsuch that zero voltage switching can be maintained for S1 and S2 over awide range of operating conditions. A soft-switching observer asdiscussed before can be used to identify whether an acceptablesoft-switching condition has been achieved. If not, the switchingfrequency can be adjusted for improving the soft switching condition.For example, the waveform of Vsw and/or the waveform Isw can indicatewhether a power switch has achieved zero voltage switching (zero-voltageturn-on). If a power switch does have soft switching, the switchingfrequency of S1 and S2 can be reduced. If both S1 and S2 have achievedsoft-switching, and at the turn-on instants of S1 and S2, Isw is toohigh or the derivative of Vsw is too high in magnitude, the switchingfrequency of S1 and S2 should be changed. By adjusting the switchingfrequency in a reasonable range, a good trade-off on efficiency may beachieved.

In some embodiments, the zero-voltage switching asymmetric half-bridgeconverter 1600 operates in a continuous conduction mode. During thecontinuous conduction mode, a current flows through D1 during the periodfrom the turn-on of S2 to the turn-off of S2. On the other hand, acurrent flows through D2 during the period from the turn-on of S1 to theturn-off of S1. The output voltage of the zero-voltage switchingasymmetric half-bridge converter 1600 can be expressed by the followingequation:

$\begin{matrix}{V_{o} = \frac{{D\left( {1 - D} \right)}{Vin}}{{K\; 1} + {D\left( {{K\; 2} - {K\; 1}} \right)}}} & (1)\end{matrix}$

where K1 is the turns ratio of the first transformer T1 (e.g., K1 isequal to the number of turns of T1 p divided by the number of turns ofT1 s); K2 is the turns ratio of the second transformer T2 (e.g., K2 isequal to the number of turns of T2 p divided by the number of turns ofT2 s). D is the duty cycle of S1.

In some embodiments, K1 is equal to K2 or D is very small. Equation (1)above can be simplified to the following equation:

$\begin{matrix}{V_{o} \approx \frac{{D\left( {1 - D} \right)}{Vin}}{K\; 1}} & (2)\end{matrix}$

Equation (2) indicates under such conditions (e.g., K1 is equal to K2 ora small duty cycle), the zero-voltage switching asymmetric half-bridgeconverter 1600 behaves like a conventional single-transformer asymmetrichalf-bridge converter, which is well known in the art, and hence is notdiscussed herein.

In some embodiments, the duty cycle D is small and K1 is much greaterthan K2 (K1>>K2). Equation (1) above can be simplified to the followingequation:

$\begin{matrix}{V_{o} \approx \frac{D\; {Vin}}{K\; 1}} & (3)\end{matrix}$

Equation (3) indicates under such conditions (e.g., D is small andK1>>K2), the zero-voltage switching asymmetric half-bridge converter1600 behaves like a conventional forward converter, which is well knownin the art, and hence is not discussed herein.

In some embodiments, the zero-voltage switching asymmetric half-bridgeconverter 1600 operates in a light load mode. In response to the lightload mode, D is reduced and the zero-voltage switching asymmetrichalf-bridge converter 1600 enters into a discontinuous conduction mode.

Under the discontinuous conduction mode, one of the transformers shownin FIG. 16 may not transfer much energy to the output. As a result, acorresponding diode coupled to its secondary winding stops conductingcurrent because the current previously flowing through the diode fallsto a level approximately equal to zero even though the correspondingprimary switch is still conducting. Under this discontinuous conductionmode, the converter behaves like a flyback converter, which is wellknown in the art, and hence is not discussed herein.

It should be noted that, during the discontinuous conduction mode, theswitching frequency of the zero-voltage switching asymmetric half-bridgeconverter 1600 can be lowered to further reduce the power losses.

In some embodiments, the zero-voltage switching asymmetric half-bridgeconverter 1600 operates in an ultra-light load mode. It should be notedthat there may be a threshold between the light load mode describedabove and the ultra-light load mode described herein. The selection ofthis threshold depends on different applications and design needs.

In response to the ultra-light load mode, the power switches S1 and S2may no longer work in the complementary mode. Instead, S2 may be turnedon just for a short time. Furthermore, it may be not necessary to applya high voltage gate drive signal to S2. For example, the turn-on time ofS2 may be limited to the conduction of the body diode of S2. Moreparticularly, S2 may be not turned on and the body diode of S2 may notconduct current. For example, the capacitance across S2 may bedischarged to some degree but not completely discharged when thedischarging current is very small.

The operation in such a mode is explained with reference to thesimulation results as shown in FIG. 18. For a short period of time, S1is turned on and operates in a normal operation mode. During the turn-ontime of S1, both the first transformer T1 and the second transformer T2are charged, and Ip is a negative current, but increases in amplitude.However, because the voltage across the primary windings, which is thevoltage across C1 is much lower than the input voltage Vin during theturn-on time of S1, neither D1 nor D2 conducts during the turn-on timeof S1. After S1 is turned off, some of the energy stored in transformersT1 and T2 is delivered to the output through D1. The body diode of S2may conduct for a short time if Ip has high enough magnitude todischarge the capacitance across S2 completely. However, in thesimulation shown in FIG. 18 below, because the energy in thetransformers T1 and T2 is low, D2 never conducts because a reversevoltage is applied to the diode D2.

In operation, the voltage across the auxiliary winding T1 a can besampled as a signal representing or indicating the output voltage Vo.When the current in D1 falls to zero or close to zero, D1 stopsconducting current, and the magnetizing inductance of T1 and T2resonates with the capacitances across S1 and S2. As a result, Vsw is ofan oscillation waveform during this time. See FIG. 18 below. The fluxbalance of T1 and T2 is maintained during this mode of operation throughthe conduction of D2 and/or the oscillation of Vsw. The conduction timeof S2, the conduction time of S1, and/or the switching frequency can beused to regulate the output voltage and/or the output power.

It should be noted that S2 doesn't need to be turned on in eachswitching cycle. S2 can be tuned on when more energy is needed.Furthermore, when S2 is turned on, the turn-on and/or the turn off of S2may be in sync with the turn-on and/or the turn-off of S1. By this way,at least one of S1 and S2 is turned on with zero voltage switching orturned on with a significantly reduced voltage across the switch.

In this ultra-light load operation mode, either S1 or S2 may not achievesoft-switching. However, because the voltage applied to the switch nothaving soft switching is low, the switching losses as well as theconduction losses may be low too. As a result, the total power loss iskept at a very low level. If necessary, the zero-voltage switchingasymmetric half-bridge converter 1600 shown in FIG. 16 may operate in aburst mode when the output power is below a suitable threshold. Theburst mode helps to further reduce the power loss at very light or noload.

The controller 1602 shown in FIG. 16 can be configured in accordancewith the different operating modes described above. As a result, thezero-voltage switching asymmetric half-bridge converter 1600 shown inFIG. 16 may achieve better performance.

In order to further reduce the cost of the zero-voltage switchingasymmetric half-bridge converter 1600 shown in FIG. 16, a primary sidecontroller may be employed. In particular, the output voltage and/orcurrent regulation circuit is placed at the primary side of thetransformers T1 and T2. When D1 conducts current, the output voltage canbe detected on the windings of T1. Likewise, when D2 conducts current,the output voltage can be detected on the windings of T2. As such, theauxiliary winding of T1 and/or the auxiliary winding of T2 can be usedto provide the information of the output voltage Vo to the controlsystem for regulating the output voltage.

As shown in FIG. 16, auxiliary windings T1 a and T2 a are both coupledto the controller 1602. The voltage information from T1 a and/or T2 acan be sampled at each switching cycle and the sampled value is used asthe feedback value of the output voltage Vo. However, in some operationmodes, D1 or D2 may not conduct, or the conduction time of D1 or D2 isnot long enough for achieving a reliable sampling of the information ofVo. Under this situation, the sampled voltages may not be used directly.Instead, the voltage waveforms from T1 a and t2 a need to be analyzed.For example, if one of the diodes doesn't conducts (which can beidentified by analyzing the waveform of this diode), the correspondingsampled voltage should not be used for Vo regulation.

In some embodiments, both of these two diodes may not conduct longenough to allow a reliable sampling of the output voltage. An offset maybe added to the previously sampled feedback value to form a new feedbackvalue. In addition, the output current information can be detected atthe primary side. The output current can be detected by sensing thecurrent Isw, sensing the current flowing through the dc link currentsuch as the voltage Vir across the current sense resistor Rs shown inFIG. 16. The sensed current information can be send into the controller1602. The controller 1602 can provide a variety of functions such ascurrent limiting and/or over current protection based on the sensedcurrent information from the primary side. The ability to regulateoutput current from the primary side may be important for someapplications such as LED lighting and the like.

FIG. 17 illustrates a variety of waveforms associated with thezero-voltage switching asymmetric half-bridge converter shown in FIG. 16in accordance with various embodiments of the present disclosure. Thehorizontal axis of FIG. 17 represents intervals of time. The unit of thehorizontal axis is microsecond. There may be seven vertical axes. Thefirst vertical axis Y1 represents the voltage (V_(SW)) across thedrain-to-source of the switch S1. The second vertical axis Y2 representsthe current (Ip) flowing through the first transformer T1. The thirdvertical axis Y3 represents the voltage (Vt2) across the primary windingof the second transformer T2. The fourth vertical axis Y4 represents thecurrent (Id2) flowing through the diode D2. The fifth vertical axis Y5represents the voltage (Vt1) across the primary winding of the firsttransformer T1. The sixth vertical axis Y6 represents the current (Id1)flowing through the diode D1. The seventh vertical axis Y7 representsthe output voltage (Vo).

In some embodiments, the input voltage of the zero-voltage switchingasymmetric half-bridge converter is about 350 V. The output voltage Vois regulated at about 19 V. The output power is about 40 W. Theoperation and the corresponding waveforms of the zero-voltage switchingasymmetric half-bridge converter have been described above with respectto FIG. 16, and hence are not discussed in further detail herein.

FIG. 18 illustrates a variety of waveforms associated with thezero-voltage switching asymmetric half-bridge converter operating in theultra-light load mode in accordance with various embodiments of thepresent disclosure. The input voltage is about 350 V and the outputvoltage is about 5 V. The ultra-light load operation and thecorresponding waveforms of the zero-voltage switching asymmetrichalf-bridge converter have been described above with respect to FIG. 16,and hence are not discussed in further detail herein.

FIG. 19 illustrates a cross sectional view of an integrated magneticstructure employed in FIG. 16 in accordance with various embodiments ofthe present disclosure. The first transformer T1 and the secondtransformer T2 shown FIG. 16 may be implemented as two individualtransformers. In alternative embodiments, these two transformers can beimplemented in an integrated magnetic structure having a single magneticcore as shown in FIG. 19.

As shown in FIG. 19, T1 is implemented around one leg of the magneticcore. T2 is implemented in another leg of the magnetic core. An air gapof exists in each of these two legs so that some energy can be stored inthe air gaps. Depending on different applications and design needs, theheight of the air gap may be selected accordingly.

A third leg or a center leg the magnetic core provides another path forthe magnetic flux in the transformers T1 and T2. An optional air gap maybe added in the third leg. This optional air gap can be used to adjustthe coupling between the transformers T1 and T2.

According to the topology shown in FIG. 16, the volts-second rating ofT1 and T2 may be reduced to a lower level in comparison with othertopologies. Such a lower volts-second rating helps to reduce the numbersof turns of T1 p and T2 p. The windings of T1 and T2 can then beimplemented on a PCB. Furthermore, the coupling between the windings ineach transformer is tightly controlled. Such a tight controlled couplinghelps to improve the control performance in PSR applications and makethe EMC design simpler.

Although embodiments of the present invention and its advantages havebeen described in detail, it should be understood that various changes,substitutions and alterations can be made herein without departing fromthe spirit and scope of the invention as defined by the appended claims.

Moreover, the scope of the present application is not intended to belimited to the particular embodiments of the process, machine,manufacture, composition of matter, means, methods and steps describedin the specification. As one of ordinary skill in the art will readilyappreciate from the disclosure of the present invention, processes,machines, manufacture, compositions of matter, means, methods, or steps,presently existing or later to be developed, that perform substantiallythe same function or achieve substantially the same result as thecorresponding embodiments described herein may be utilized according tothe present invention. Accordingly, the appended claims are intended toinclude within their scope such processes, machines, manufacture,compositions of matter, means, methods, or steps.

What is claimed is:
 1. A system comprising: a primary switch networkcoupled to a power source, wherein the primary switch network comprisesa plurality of power switches; a primary resonant tank coupled to theplurality of power switches, wherein a resonant capacitor of the primaryresonant tank is formed by a first variable capacitance network, andwherein the first variable capacitance network is modulated to improvesoft switching of the plurality of power switches through reducing avoltage level and a current level of a switch at a turn-on instant; anda primary coil coupled to the primary resonant tank.
 2. The system ofclaim 1, wherein: the first variable capacitance network comprises afirst capacitor and a diode connected in series, and a plurality ofswitch-capacitor networks connected in parallel with the diode, andwherein switches in the plurality of switch-capacitor networks areconfigured to change a capacitance of the first variable capacitancenetwork.
 3. The system of claim 1, further comprising: a secondary coilmagnetically coupled to the primary coil; and a secondary resonant tankhaving a second variable capacitance network coupled to an output,wherein a capacitance of the second variable capacitance network ismodulated to regulate a voltage or power at the output during normaloperation.
 4. The system of claim 3, wherein: the capacitance of one ofthe variable capacitance networks is adjusted to protect the systemduring a fault.
 5. The system of claim 3, wherein: the capacitance ofone of the variable capacitance networks is adjusted gradually toimplement a soft start process.
 6. The system of claim 1, furthercomprising: a soft switching observer having a first input receiving afeedback signal representing a current flowing through the primaryswitch network; and a control and protection unit configured to reduce aswitching current at a switching instant of the switch and generate gatedrive signals for the plurality of power switches at a first output ofthe control and protection unit and a control signal for adjusting acapacitance of the first variable capacitance network at a second outputof the control and protection unit.
 7. The system of claim 6, wherein:the feedback signal is generated from a voltage across the switch. 8.The system of claim 1, further comprising: an EMI filter comprising aplurality of resonant circuits, wherein each resonant circuit has acapacitor and an inductor, and wherein the capacitor is trimmed so thata resonant frequency of a resonant circuit comprising the trimmedcapacitor is approximately equal to a predetermined value.
 9. A methodfor a power apparatus comprising: detecting a signal representing acurrent level at a turn-on transition of a power switch of a primaryswitch network, wherein the primary switch network is coupled to aprimary resonant tank having a resonant capacitor formed by a firstvariable capacitance network comprising a plurality of switch-capacitornetworks; and modulating a capacitance of the first variable capacitancenetwork to improve soft switching of the power switch through reducing avoltage level and a current level at the turn-on transition of the powerswitch.
 10. The method of claim 9, further comprising: graduallyadjusting a capacitance of the first variable capacitance network toimplement a soft start of the power apparatus.
 11. The method of claim9, wherein: the primary switch network and the primary resonant tank aremagnetically coupled to a secondary side comprising a secondary resonanttank, wherein the secondary resonant tank comprises a secondary resonantcapacitor formed by a second variable capacitance network.
 12. Themethod of claim 11, further comprising: adjusting a capacitance of thesecond variable capacitance network so that an output voltage or outputcurrent of the secondary side is regulated at a predetermined level. 13.The method of claim 11, further comprising: modulating the capacitanceof one of the variable capacitance networks; and establishing acommunication channel through sensing a signal change during the step ofmodulating the capacitance of one of the variable capacitance networks.14. The method of claim 11, further comprising: adjusting thecapacitance of the second variable capacitance network gradually toachieve a soft start process or a soft stop process of at an output ofthe secondary side.
 15. The method of claim 11, further comprising:adjusting the capacitance of one of the variable capacitance networks toprotect the power apparatus during abnormal operating conditions.
 16. Amethod comprising: providing a wireless power transfer system comprisinga transmitter magnetically coupled to a first receiver, wherein: thetransmitter comprises a power amplifier coupled to an input powersource, a transmitter resonant tank comprising a first variablecapacitance network and a transmitter coil coupled to the transmitterresonant tank; and the first receiver comprises a first receiverresonant tank comprising a first receiver variable capacitance networkand a first receiver coil coupled to the first receiver resonant tank;and modulating a capacitance of the first variable capacitance networkso that switches of the power amplifier achieve an improved softswitching condition.
 17. The method of claim 16, further comprising:detecting a current flowing through the transmitter; and modulating thecapacitance of the first variable capacitance network based upon thedetected current to maintain a current in the transmitter at apredetermined level at a switching instant.
 18. The method of claim 17,wherein: a control speed of the step of modulating the capacitance ofthe first variable capacitance network is based upon the detectedcurrent.
 19. The method of claim 16, further comprising: coupling asecond receiver to the transmitter, wherein the second receivercomprises a second receiver resonant tank comprising a second receivervariable capacitance network and a second receiver coil coupled to thesecond receiver resonant tank, and wherein the step of coupling thesecond receiver to the transmitter is implemented in a smooth manner byadjusting a capacitance of the second receiver variable capacitancenetwork.
 20. The method of claim 19, further comprising: decoupling thesecond receiver from the transmitter in a smooth manner by adjusting thecapacitance of the second receiver variable capacitance network.